This invention relates to a high temperature superconductor (HTS) tunable filter. More particularly, this invention relates to an HTS filter tunable by actuating a magnetic driver.
The need for a high-quality factor (Q), low insertion loss tunable filter pervades a wide range of microwave and RF applications, in both the military, e.g., RADAR, communications and Electronic Intelligence (ELINT), and the commercial fields such as in various communications applications, including cellular. Placing a sharply defined bandpass filter directly at the receiver antenna input will often eliminate various adverse effects resulting from strong interfering signals at frequencies near the desired signal frequency in such applications. Because of the location of the filter at the receiver antenna input, the insertion loss must be very low to not degrade the noise figure. In most filter technologies, achieving a low insertion loss requires a corresponding compromise in filter steepness or selectivity. In the present invention, the extremely low loss property of high-temperature superconductor (HTS) filter elements provides an attractive solution, achieving a very low insertion loss yet simultaneously allowing a high selectivity/steepness bandpass definition.
In many applications, particularly where frequency hopping is used, a receiver filter must be tunable to either select a desired frequency or to trap an interfering signal frequency. The vast majority of lumped element tunable filters have used varactor diodes. Such a design amounts to using a tunable capacitor because varactor diodes, by changing the reverse bias voltage, vary the depletion thickness and hence the P-N junction capacitance. While varactors are simple and robust, they have limited Q""s, and suffer from the problem that the linear process that tunes them extends all of the way to the signal frequency, so that high-amplitude signals create, through the resulting nonlinearities, undesirable intermodulation products and other problems.
Consider the case of a conventional varactor diode. In a varactor, the motion of electrons accomplishes the tuning itself. As the reverse bias voltage (Vr) on the junction of the varactor is changed, then in accordance with Poisson""s Equation, the width of the P-N junction depletion region changes, which alters the junction capacitance (Cj). Because the tuning mechanism of varactors is electronic, the tuning speed is extremely fast. Unfortunately, this also leads to a serious associated disadvantage: limited dynamic range. Because the Cj (Vr) relationship is nearly instantaneous in response, extending to changes in Vr at the signal frequency itself, and the input signal (frequently in a resonantly magnified form) appears as a component of the junction bias voltage Vr, the input signal itself parametrically modulates the junction capacitance. If the signal amplitude across the varactor is very small in comparison to the dc bias, the effect is not too serious. Unfortunately, for high signal amplitudes, this parametric modulation of the capacitance can produce severe cross-modulation (IM) effects between different signals, as well as harmonic generation and other undesirable effects. While these signal-frequency varactor capacitance variations are the basis of useful devices such as parametric amplifiers, subharmonic oscillators, frequency multipliers, and many other useful microwave circuits, in the signal paths of conventional receivers they are an anathema. This inherent intermodulation or dynamic range problem will presumably extend to xe2x80x9ctunable materialsxe2x80x9d, such as ferroelectrics or other materials in which the change of dielectric constant (∈r) with applied electric field (E) is exploited to tune a circuit. As long as the ∈r (E) relationship applies out to the signal frequency, then the presence of the signal as a component of E will lead to the same intermodulation problems that the varactors have.
In addition to the intermodulation/dynamic range problems of varactors, these conventional tuning devices also have serious limitations in Q, or tuning selectivity. Because the varactors operate by varying the depletion region width of a P-N junction, at lower reverse bias voltages (higher capacitances), there is a substantial amount of undepleted moderately-doped semiconductor material between the contacts and the P-N junction that offers significant series resistance (Rac) to ac current flow. Since the Q of a varactor having junction capacitance Cj and series resistance Rac at an input signal frequency f is given by Q=1/(2 f Cj Rac), the varactor Q values are limited, particularly at higher frequencies. For example, a typical commercial varactor might have Cj=2.35 pF with Rac, =1.0xcexa9 at Vr=xe2x88x924V, or Cj=1.70 pF with Rac=0.82xcexa9 at Vr=xe2x88x9210V, corresponding to Q values at f=1.0GHz of Q=68 at Vr=xe2x88x924V or Q=114 at Vr=xe2x88x9210V (or f=10.0 GHz values of Q=6.8 and Q=11.4, respectively). Considering that an interesting X-band (f=10 GHz) RADAR application might want a bandwidth of xcex94F=20 MHz (FWHM), corresponding to a Q=f/xcex94F=500 quality factor, we see that available varactors have inadequate Q (too much loss) to meet such requirements. While the mechanisms are different, this will very likely apply to the use of ferroelectrics or other xe2x80x9ctunable materials.xe2x80x9d A general characteristic of materials which exhibit the field-dependent dielectric constant nonlinearities (that makes them tunable) is that they exhibit substantial values of the imaginary part of the dielectric constant (or equivalently, loss tangent). This makes it unlikely that, as in varactors, these xe2x80x9ctunable materialsxe2x80x9d will be capable of achieving high Q""s, particularly at high signal frequencies.
An additional problem with both varactors and xe2x80x9ctunable materialsxe2x80x9d for circuits with high values of Q is that these are basically two-terminal devices; that is, the dc tuning voltage must be applied between the same two electrodes to which the signal voltage is applied. The standard technique is to apply the dc tuning bias through a xe2x80x9cbias teexe2x80x9d-like circuit designed to represent a high reactive impedance to the signal frequency to prevent loss of signal power out the bias port (as this loss would effectively reduce the Q). However, while the design of bias circuits that limit the loss of energy to a percent, or a fraction of a percent of the resonator energy is not difficult, even losses of a fraction of a percent are not nearly good enough for very high Q circuits (e.g., Q""s in the 103 to  greater than 105 range, as achievable with HTS resonators). It would be much easier to design such very high Q circuits using three-terminal, or preferably 4-terminal (two-port) variable capacitors in which the tuning voltage is applied to a completely different pair of electrodes from those across which the input signal voltage is applied (with an inherent high degree of isolation between the signal and bias ports).
One new form of variable capacitor that avoids the intermodulation/dynamic range problems of varactors or xe2x80x9ctunable materialsxe2x80x9d approaches is the microelectromechanical (MEMS) variable capacitor. A number of MEMS variable capacitor device structures have been proposed, including elaborate lateral-motion interdigitated electrode capacitor structures. In the simple vertical motion, parallel plate form of this device, a thin layer of dielectric separating normal metal plates (or a normal metal plate from very heavily doped silicon) is etched out in processing to leave a very narrow gap between the plates. The thin top plate is suspended on four highly compliant thin beams which terminate on posts (regions under which the spacer dielectric has not been removed). The device is ordinarily operated in an evacuated package to allow substantial voltages to be applied across the narrow gap between plates without air breakdown (and to eliminate air effects on the motion of the plate and noise). When a dc tuning voltage is applied between the plates, the small electrostatic attractive force, due to the high compliance of the support beams, causes substantial deflection of the movable plate toward the fixed plate or substrate, increasing the capacitance.
Because the change of capacitance, at least in the metal-to-metal plate version of the MEMS variable capacitor, is due entirely to mechanical motion of the plate (as opposed to xe2x80x9cinstantaneousxe2x80x9d electronic motion effects as in varactors or xe2x80x9ctunable materialsxe2x80x9d), the frequency response is limited by the plate mass to far below signal frequencies of interest. Consequently, these MEMS devices will be free of measurable intermodulation or harmonic distortion effects, or other dynamic range problems (up to the point where the combination of bias plus signal voltage across the narrow gap between plates begins to lead to nonlinear current leakage or breakdown effects).
In addition to their freedom from intermodulation/dynamic range problems, normal metal plate MEMS variable capacitor structures offer the potential for substantially lower losses and higher Q""s. While the simple parallel plate MEMS structure has a Q problem due to the skin effect resistance, Rac, of the long narrow metal leads down the compliant beams supporting the movable plate, an alternative structure is possible which avoids this problem. If the top (movable) plate is made electrically xe2x80x9cfloatingxe2x80x9d (from a signal standpoint, it would still have a dc bias lead on it), and the fixed bottom plate split into two equal parts, these two split plates can be used as the signal leads to the MEMS variable capacitor. (The capacitance value is halved, of course, but the tuning range is preserved.) In this xe2x80x9cfloating platexe2x80x9d configuration, passage of ac current through the long narrow beam leads is avoided, allowing fairly high values of Q to be achieved, even with normal metal plates.
While this conventional MEMS variable capacitor structure is capable of improved Q""s and avoids the intermodulation problems of varactors and xe2x80x9ctunable materialsxe2x80x9d, it has some potential problems of its own. For example, the electrostatic force attracting the two plates is quite weak, except at extremely short range. The electrostatic force Fe between two parallel plates each of area A with a voltage difference V and a gap separation z is given by
Fe=xe2x88x92(∈0A/2)(V/z)2xe2x80x83xe2x80x83(Eq. 1)
where ∈0=8.854xc3x9710xe2x88x9212 Farad/Meter (F/m) is the permittivity of a vacuum. The extremely rapid falloff of force as the separation gap is increased (as 1/z2) makes the useful tuning range of electrostatic drivers quite small. In this parallel-plate MEMS capacitor configuration, if a linear spring provides the restoring force between the plates, when the bias voltage is increased such that the gap separation has dropped to 1/3 of the separation at zero bias, the plate motion becomes unstable and the plates snap together. This limits the useful tuning range to less than 3:1 in capacitance, or less than 1.732:1 in frequency. Further, the short-range nature of the electrostatic force makes its use in variable-inductance tuning even more problematic because of the requirement for very narrow gaps (to give reasonable levels of force at reasonable drive voltages), since much larger gaps (e.g., hundreds of microns) are desirable in devices having such variable-inductance tuning.
The short-range nature of the electrostatic force is illustrated by the following example. In a parallel-plate capacitor having a voltage of 100 volts (which is actually an unreasonably high voltage level given the trends toward low voltage electronics) and a gap separation of 1.0 xcexcmeter (xcexcm), the electrostatic force (divided by the area of the plates) is 4.514 grams/centimeter2, a reasonable force. Increasing the gap to 10 xcexcm at the same voltage produces the minuscule attractive force of 0.04514 grams/centimeter2. On the other hand, decreasing the gap to 0.1 xcexcm at the same voltage produces the robust attractive force of 451.43 grams/centimeter2, corresponding to an electric field strength of 107 V/cm. Although coating the plates with a thin dielectric and allowing progressive contact of thin curved (stress-bent) layers with a fixed electrode as voltage is increased may counteract the short-range effect of this electrostatic force (and with proper drive plate shaping, extend the tuning range in capacitance beyond 3:1), triboelectric (i.e., charging due to friction) and charge transfer effects under the high field condition tend to give significant hysteresis in the capacitance-voltage (C-V) characteristics of these xe2x80x9cwindow shadexe2x80x9d MEMS devices.
In addition, there are other potential problems in conventional MEMS devices. For example, in many system applications for tunable filters, requirements for precise phase make it essential that the selected frequency be very stable and reproducible. Consider a resonator or narrowband filter having a center frequency Fo and a xe2x88x923 dB bandwidth xcex94F given from its (loaded) quality factor Qo by the equation
xcex94F=Fo/Qoxe2x80x83xe2x80x83(Eq. 2)
Note that as the frequency is changed from (Foxe2x88x92xcex94F/2) through Fo to (Fo+xcex94F/2), the phase changes quite dramatically from +45xc2x0 to 0xc2x0 to xe2x88x9245xc2x0. For a signal frequency f near Fo, the phase in a single resonator may be approximated by
Phase (xc2x0)≈2Qo(180xc2x0/xcfx80)[1xe2x88x92(f/Fo)]xe2x80x83xe2x80x83(Eq. 3)
(for a single resonator, or Nr times this value for a filter having Nr resonators at Fo). Hence, if the allowable phase uncertainty at a given frequency f is denoted by xcex94Phase (xc2x0), then the allowable error in the resonator center frequency xcex94Fo, near resonance will be
xcex94Fo/f=xcex94Phase(xc2x0)/[2Qo(180xc2x0/xcfx80)]=(0.0087266/Qo)xcex94Phase(xc2x0)xe2x80x83xe2x80x83(Eq. 4)
For example, for a 1.0xc2x0 degree phase error with a loaded Qo=500, the resonator frequency repeatability, xcex94Fo/f, must be less than or equal to 0.00175 % (for a single resonator, or 1/Nr times this value for a number Nr of resonators). This means that for such phase sensitive applications, the tunable elements must achieve levels of repeatability, hysteresis and continuity that appear difficult to achieve in ferroelectric piezoelectric actuators, let alone xe2x80x9cwindow shadexe2x80x9d electrostatic MEMS devices.
Therefore, there is a need in the art for new driver structures for varying the properties of MEMS-like HTS capacitors or inductors, or more complex distributed resonator structures having transmission line-like qualities. The resulting variable capacitors, inductors, or other tunable elements may be incorporated into tunable filters or other circuits.
In one innovative aspect, the present invention comprises a circuit wherein the electronic properties of the circuit are varied by altering the current through a magnetic actuator. The circuit includes a fixed substrate and a movable substrate wherein the magnetic actuator alters the position of the movable substrate with respect to the fixed substrate. The magnetic actuator comprises a magnetic driver having a continuous strip of HTS material on an upper surface of the fixed substrate. Note that as used herein, a xe2x80x9ccontinuous strip of HTS materialxe2x80x9d will include within its scope a strip of HTS material that may be interrupted by segments of non-HTS materials such as normal metals used in overcrossings. A lower surface of the movable substrate opposes the upper surface of the fixed substrate. On the lower surface, the magnetic actuator includes an HTS reaction plate substantially overlapping the magnetic driver whereby a tuning current flowing through the continuous strip of HTS material produces a repulsive force between the magnetic driver and the HTS reaction plate.
In one embodiment, the circuit includes a split-plate variable capacitor. The variable capacitor comprises a first capacitor plate and a second capacitor plate on the upper surface of the fixed substrate and a floating capacitor plate on the lower surface of the movable substrate that substantially overlaps the first and second capacitor plates wherein the first and second capacitor plates opposing the floating capacitor plate define a gap of the variable capacitor. As current flows through the magnetic driver, the repulsive force induced between the magnetic driver and the HTS reaction plate changes the capacitor gap, thereby varying the capacitance of the variable capacitor.
In another embodiment of the invention, the circuit includes a variable inductor. The variable inductor comprises an HTS inductor on the upper surface of the fixed substrate and an HTS inductance suppression plate on the lower surface of the movable substrate that substantially overlaps the HTS inductor.
A restoring force that opposes the force produced by the magnetic actuator may be provided by a first and a second membrane attached to a first and second end of the movable substrate, respectively. The first membrane connects the first end of the movable substrate to a first post. on the upper surface of the fixed substrate, the first post being laterally disposed to the first end of the movable substrate. Similarly, the second membrane connects the second end of the movable substrate to a second post on the upper surface of the fixed substrate, the second post being laterally disposed to the second end of the movable substrate.
The force generated by the magnetic actuator that moves the movable substrate with respect to the fixed substrate may be either a xe2x80x9cpushxe2x80x9d (repulsion only) or a xe2x80x9cpush-pullxe2x80x9d (repulsion/attraction) type force. In embodiments of the invention in which the HTS reaction plate has neither any trapped magnetic flux nor any permanent magnets, the magnetic actuator is a push magnetic actuator. HTS reaction plates for a push magnetic actuator are preferably solid plates. In a push-pull magnetic actuator, the actuator may include trapped circulating supercurrents within the HTS reaction plate to generate an attractive magnetic force that interact with the driver current in such a way as to produce, for one direction of driver current, an enhanced repulsive force, while for driver currents within a certain range of magnitude in the opposite direction, an attractive force is created between the driver and this xe2x80x9cpoledxe2x80x9d reaction plate. This attractive magnetic force would, if otherwise unopposed by application of spring-like mechanical restoring force, tend to draw the movable substrate towards the fixed substrate. Suitable HTS reaction plates for a push-pull magnetic actuator preferably comprise at least one concentric closed loop of HTS material and may conveniently be referred to as a xe2x80x9cpoledxe2x80x9d HTS reaction plate, in analogy with terminology used for ferromagnetic or ferroelectric devices. Circulating supercurrents that are held within the xe2x80x9cpoledxe2x80x9d HTS reaction plate generate a magnetic flux that has a component parallel to the plate. This field component may produce an attractive xe2x80x9cpullxe2x80x9d force between the reaction plate and the driver coil if the driver current is in the correct polarity and magnitude, thus providing the xe2x80x9cpullxe2x80x9d within a push-pull magnetic actuator. Alternatively, conventional permanent magnet material poled to attract the magnetic driver could be incorporated into the movable substrate adjacent the HTS reaction plate to provide a push-pull magnetic actuator.
The present invention also includes methods of inducing the circulating supercurrents within a xe2x80x9cpoledxe2x80x9d HTS reaction plate of a push-pull magnetic actuator. In one method, the magnetic driver is cooled below its critical temperature while the HTS reaction plate is above its critical temperature and the HTS reaction plate and the magnetic driver are in close proximity. A drive current is then induced in the magnetic driver while the HTS reaction plate is cooled below its critical temperature, thereby inducing the circulating supercurrents within the continuous strip of HST material to xe2x80x9cpolexe2x80x9d the xe2x80x9cpoledxe2x80x9d HTS reaction plate. To assist cooling the magnetic driver below its critical temperature while the magnetic driver is in close proximity to a HTS reaction plate above its critical temperature, the magnetic driver may be constructed from HTS material that has a higher critical temperature than the HTS material used to construct the HTS reaction plate. Alternatively, both the magnetic driver and the HTS reaction plate may be brought below their critical temperatures. Then, a heat source above an upper surface of the movable substrate may generate radiant energy to briefly raise the HTS reaction plate above its critical temperature without raising the magnetic driver above its critical temperature while a drive current is applied to the magnetic driver coil.
An alternative method does not require the application of a drive current through the magnetic driver. Instead, both the magnetic driver and the HTS reaction plate are cooled below their critical temperatures. Then, a high intensity pulsed magnetic field aligned normally to the lower surface of the movable substrate would be applied to induce the circulating supercurrents within the continuous strip of HTS material to (xe2x80x9cpolexe2x80x9d) the xe2x80x9cpoledxe2x80x9d push-pull driver reaction plate.
In an another embodiment of the invention, opposing push magnetic actuators are used to provide a xe2x80x9cpush-pullxe2x80x9d operation despite the absence of a push-pull magnetic actuator. In one embodiment, the movable substrate lies between opposing surfaces of the fixed substrate wherein the opposing surfaces of the fixed substrate are spaced apart a distance greater than the thickness of the movable substrate, thereby allowing translational movement of the movable substrate between the opposing surfaces. A first magnetic actuator comprises a magnetic driver on one of the opposing surfaces of the fixed substrate. A first HTS reaction plate on the surface of the movable substrate opposing the first magnetic driver substantially overlaps the first magnetic driver. A second magnetic actuator comprises a magnetic driver on the other of the opposing surfaces of the fixed substrate. A second HTS reaction plate on the surface of the movable substrate opposing the second magnetic driver substantially overlaps the second magnetic driver, whereby the second and first magnetic actuators produce opposing forces on the movable substrate. Alternatively, a single HTS reaction plate on one of the sides of the movable substrate may be used to generate the repulsive reaction forces from both the first magnetic driver and the second magnetic driver.
In an another embodiment, the movable substrate is suspended on a torsionally compliant fiber or band. The torsion fiber attaches to and extends across the upper surface of the movable substrate. Preferably, the torsion fiber is positioned on a centerline of the movable substrate such that, absent additional forces, the lower surface of the suspended movable substrate is parallel to the upper surface of the fixed substrate. The torsion fiber may be attached to posts on the fixed substrate that are laterally disposed to the movable substrate. A first and a second magnetic actuator are located on opposite sides of the torsion fiber. Rotational motion of the torsionally suspended movable substrate is induced in one direction when current is passed through the driver coil on one side of the torsion fiber axis, and in the opposite direction when the current is passed through the opposing driver on the other side of the rotational axis. In a preferred embodiment, to allow a greater tuning range, the movable substrate comprises a first and a second planar portion attached to each other in a dihedral configuration, the torsion fiber axis being located near the apex of the dihedral angle. This dihedral angle allows the rotational axis of the movable substrate to be placed very close to the fixed substrate, while still permitting rotation of the movable substrate by an angle slightly greater than the dihedral angle without either of the sides of the movable substrate striking the fixed substrate. The dihedral configuration allows a planar portion of the movable substrate to go from a tuning position parallel to, and in very close proximity to, the fixed substrate, to a rotated position in which the end of the planar portion is a comparatively large distance from the fixed substrate (and angled away from it by the dihedral angle). This enables a very large tuning range to be achieved in either capacitive or inductive tuning (or combinations of these in complex resonator structures). In an alternate embodiment, the movable substrate comprises a first planar portion and a second planar portion wherein the first and second planar portions are joined with a lap joint. The torsion fiber would attach to the movable substrate adjacent the lap joint.
While the use of a rotationally compliant torsion fiber or band suspension has been described here, a number of different mechanical means to constrain the position of the axis of rotation of the movable substrate to obtain very low friction and backlash (hysteresis), and nearly-pure rotational motion of the movable substrate could be utilized in this embodiment of the invention. These include a fulcrum or knife edge on the movable substrate working against a flat surface, or a groove or other suitable positioning structure on the fixed substrate, a fulcrum or knife edge on the fixed substrate working against a flat surface, or a groove or other suitable positioning structure on the movable substrate, or the combination of one of these with a torsion fiber or band to assist in maintaining proper positioning of the movable substrate and its rotational axis.